1. Field of the Invention
This invention relates to a PWM (Pulse Width Modulation) inverter system for producing a desired multi-phase ac output from a dc input by on-off controlling a switching element such as, for example, a transistor, and more particularly to production of a switching signal, that is, a PWM signal for causing an on-off operation.
2. Prior Art
Conventionally, a typical method of producing an PWM signal involves comparison of a sinusoidal wave signal which is each phase voltage instruction with a triangular wave signal which is a modulating wave. The principle of the method is described, for example, taking a single phase inverter of a full bridge as an example in "Inverter Application Manual" p.p. 28 to 34, published on Sept. 7, 1985 (hereinafter referred to as prior art 1). Of late, a method is already known wherein a PWM signal of a multi-phase inverter is treated as composition of space voltage vectors. According to the method, on-off times of individual switching elements are controlled such that a time average vector for a unit time may coincide with an instruction value of a desired instantaneous voltage vector (Japanese patent application No. 59-251001, which will be hereinafter referred to as prior art 2).
FIG. 1 shows a circuit construction of a single phase inverter of the prior art 1. In FIG. 1, reference symbols S.sub.1 to S.sub.4 denote each a switching element such as a transistor, and FIG. 2 illustrates a process of producing switching signals (on-off signals) to be coupled to the switching elements. Referring to FIG. 2, (a) shows a relationship between a sinusoidal wave signal and a triangular wave which is a modulating wave, and a sinusoidal wave signal when the magnitude thereof is great, that is, upon outputting of a high voltage, is indicated by a solid line while a sinusoidal wave upon outputting of a low voltage is indicated by a chain line. Meanwhile, (b) of FIG. 2 shows on-off signals of the individual switching elements upon outputting of the high voltage while (c) of FIG. 2 shows on-off signals upon outputting of the low voltage.
Now, the prior art 2 will be described with reference to FIGS. 3 to 8.
Referring to FIG. 3, a main circuit 1 of a 3-phase inverter of the PWM control method (hereinafter referred to simply as inverter) includes switching elements S.sub.aP, S.sub.bP, S.sub.cP, S.sub.aN, S.sub.bN and S.sub.cN such as transistors connected in a bridge circuit. The inverter 1 is connected to primary windings 2 of a 3-phase induction motor serving as a load to the inverter 1, and also to a firing signal generating circuit 3 for producing a firing signal for turning the switching elements on and off.
The voltage vectors V which are produced when a voltage to be applied to the 3-phase load shown in FIG. 3 is supplied from the inverter 1 only include such discrete voltage vectors V.sub.0 to V.sub.7 corresponding to on-off states of the switching elements of the inverter 1 as seen in FIG. 4. It can be considered that the PWM inverter produces equivalent continuous voltage vectors by switching such discrete voltage vectors at a high speed. Symbols (0, 0, 0), (1, 0, 0), . . . , and (1, 1, 1) in FIG. 4 represent on-off states of switching element pairs S.sub.a (S.sub.aPLL , S.sub.aN), S.sub.b (S.sub.bP, S.sub.bN) and S.sub.c (S.sub.cP, S.sub.cN), wherein symbol 1 represents that the switching element indicated by the suffix P is on and the switching elements indicated by the suffix N is off while symbol 0 represents an individually reverse state. However, (1, 1, 1) and (0, 0, 0) both represent a short-circuited condition wherein the load terminals are short-circuited by the switching elements, and the voltage vectors then are the zero vector having a magnitude equal to zero.
The prior art 2 describes such combinations and producing times of the voltage vectors V.sub.0 to V.sub.7 that an average voltage vector in a unit time T.sub.1 of discrete voltage vectors may coincide with a desired arbitrary instantaneous voltage vector. When, for example, a voltage vector V* of a magnitude .vertline.V*.vertline. is to be produced in a region defined by and between the vectors V.sub.1 and V.sub.2 in FIG. 4, the voltage vectors V.sub.1, V.sub.2 and 0 are produced with producing times T.sub.a, T.sub.b and T.sub.o which are defined respectively by the following expressions: ##EQU1## where T.sub.a +T.sub.b +T.sub.o =T.sub.I, and .theta.', is an angle of V* from V.sub.1 and K is a coefficient including an input dc voltage Ed.
The order of production of discrete voltage vectors V.sub.1, V.sub.2 and 0 are not limited at all from the principle of the average in time. In particular, an average voltage vector becomes identical whether the vectors are changed over in an order of V.sub.1 .fwdarw.V.sub.2 .fwdarw.0 or in another order of 0.fwdarw.V.sub.2 .fwdarw.V.sub.1 with the times provided at the time T.sub.I by the equations (1) above, and accordingly an identical average vector is obtained in an arbitrary changing over order. In this instance, however, transitions of the on-off states (switching transitions) of the switching elements are different from each other. FIG. 5 shows an example of switching transitions of the individual switching element pairs S.sub.a, S.sub.b and S.sub.c when the following two changing over orders are employed in two successive unit times 2.multidot.T.sub.I. EQU 0.fwdarw.V.sub.1 &lt;V.sub.2 .fwdarw.0.fwdarw.V.sub.2 .fwdarw.V.sub.1 (a) EQU V.sub.1 .fwdarw.V.sub.2 .fwdarw.0.fwdarw.0.fwdarw.V.sub.2 .fwdarw.V.sub.1 (b)
Here, the 0 vector can take two switching states of (0, 0, 0) and (1, 1, 1). However, in case such switching operations as put in parentheses in (a) and (b) of FIG. 5, the switching element pair S.sub.b makes twice switching operations both in the case of (a) and (b). Accordingly, it is not preferable in that the loss by the elements or the loss by the driving circuit involved in the switching operations will increase or will concentrate upon a particular phase. Further, comparison between the switching transition views of (a) and (b) of FIG. 5 reveals that while each phase switching element makes, in the case of (a), a switching operation once within the period of 2.multidot.T.sub.I, the switching element pair S.sub.a maintains the state of 1 in the case of (b) and is not required to make a switching operation so that averaging is attained by switching of each of the switching element pairs S.sub.b and S.sub.c once. Accordingly, the case of (b) of FIG. 5 is advantageous from the point of view of the loss described above.
Thus, switching signlas which cause such switching transitions required for switching operations for all the three phases as in the case of the switching transition view (a) of FIG. 5 will be hereinafter referred to as 3-phase demodulation switching signals and such 3-phase modulation as described just above will be hereinafter referred to only as 3-phase modulation while switching signals which do not require a switching operation for one phase as in the case of the switching transition view (b) of FIG. 5 will be hereinafter referred to as 2-phase modulation switching signals and such 2-phase modulation will be hereinafter referred to only as 2-phase modulation. Thus, when the triangular wave comparison method of the prior art 1 is applied to a 3-phase inverter, the switching transition then takes the transition pattern of (a) of FIG. 5 and makes 3-phase modulation. As well known in the art, such 2-phase modulation as described above is recently more advantageous than comparison of a sinusoidal wave signal with a triangular wave with respect to voltage utilizing rate (ratio of a maximum ac output voltage to a dc voltage input) and effective value of high frequency components.
If the switching elements of each of the switching element pairs S.sub.a (S.sub.aP, S.sub.aN), S.sub.b (S.sub.bP, S.sub.bN) and S.sub.c (S.sub.cP, S.sub.cN) are turned on at the same time, then the dc voltage Ed is short-circuited to cause breakdown of the switching elements. Accordingly, the switching elements of each switching element pair must be controlled such that one of them may assume the on state when the other assumes the off state. In this instance, however, when a switching element is turned from the on state to the off state, there is some delay which arises in the drive circuit for the switching element or in the switching element itself. Accordingly, a short-circuiting preventing circuit is required for delaying a signal for turning on the other element at a time other than its regular timing. FIG. 6 shows an example of such a short-circuiting preventing circuit for one phase. The short-circuiting preventing circuit of FIG. 6 operates to delay the turning on signal for a switching element connected thereto by a predetermined interval of time T.sub.d by means of an on delay element 71. In FIG. 6, symbols S.sub.x represents a switching signal, S.sub.xP * a positive side switching signal, S.sub.xN * a negative side switching signal, S.sub.xP a positive side turning on signal, and S.sub.xN a negative side turning on signal. Accordingly, the turning on signal for each of the switching elements is modified by such a non-linear element as described above.
In the case of the 2-phase modulation, there is a problem that the influence of the modification is so great that the distortion rate of the inverter output voltage becomes very high in a low voltage region wherein the rate of the producing time of the zero vector 0 is high. Description of this will be given below. Referringto the switching transition view for the 2-phase modulation shown in (b) of FIG. 5, as the producing time T.sub.0 of the zero vector 0 increases and the rate of the times T.sub.a and T.sub.b decreases, the times when the switching elements S.sub.b and S.sub.c continue the state of 0 decrease. Signals of the short-circuiting preventing circuit when such switching signals (PWM signals) are requested are illustrated in FIG. 7 wherein the turning on signal S.sub.xN of the switching element on the negative side disappears due to the short-circuiting preventing period T.sub.d and always presents the off state. Further, since all of the turning on signals having a pulse width smaller than the short-circuiting preventing period T.sub.d apparently disappear, the distortion rate of the output voltage increases remarkably due to the fact that the desired output voltage is low because it is impossible to control the output voltage by fine adjustment of the pulse width then. Due to this influence, particularly when power is supplied to a reactance load such as an induction motor, the distortion of the load current is so great that the induction motor cannot be operated stably because the frequency and the voltage have a proportional relationship and accordingly it is in a region wherein the reactance value is very small.
Further, even if a current minor loop for detecting a load current to change a voltage instruction value to cause the deviation from a current instruction value to be reduced to zero is added, disappearance of the pulses will act as a kind of blind sector. Accordingly, an effect of improvement in waveform by the minor loop cannot be anticipated and an unstable element of the system may be provided to increase the current distortion.